Volume 3 No.3, Fall 1999

ISSN# 1523-9926

Applications of Switched-Capacitor Circuits in Active Filters and Instrumentation Amplifiers

 Dr. William R. Grisé
w.grise@morehead-st.edu
Department of IET
Morehead State University


ABSTRACT

 This paper will explain the basic concepts underlying the operation of the switched capacitor, as well as the use of switched-capacitors to realize compact and versatile circuits already familiar to the undergraduate student of electronics. One set of example circuits include easily tunable active filters; specific examples of filter designs that incorporate switched-capacitors will be developed, and the use of a commercially available switched-capacitor integrated circuit, the MF10, to implement the designs will be shown. Another example circuit is an instrumentation amplifier that is more compact and has a higher CMRR than the conventional realization. Linear Technology's LTC1043 serves as the vehicle for this circuit. By demonstrating the utility of the modern switched-capacitor IC in these two important electronic functions, it is hoped that instructors and students in engineering technology will include the study of the switched-capacitor in advanced electronics courses.

 

I. INTRODUCTION

This paper aims to show how the switched-capacitor concept can be used to realize a wide variety of active filters that have the advantages of compactness and tunability.  In particular, the explanations and design examples presented here will use mathematical tools familiar to the electronics technology and engineering undergraduate student. We will not use the Z-transform, which is the rigorously correct tool for analyzing sampled-data waveforms.  

The paper will present the following topics. First, the basic ideas behind the use of the switched-capacitor to replace resistors in active filter circuits will be explained.  Second, the use of the switched-capacitor to implement lossless, lossy, and differential integrators, which are the backbone of many switched-capacitor filter circuits, especially those based on National Semiconductor's MF10 IC [1].  Third, example designs of active filters using the MF10 will be presented. 

Before detailing the operation of switched-capacitor circuits, it will be useful to understand the motivation behind, and applications of, these circuits. Basically, switched-capacitor techniques have been developed in order to allow for the integration on a single silicon chip of both digital and analog functions. Because very large scale integrated (VLSI) circuits rely on MOS transistors and pico-farad range MOS capacitors, any realization of analog circuits on a chip will have to use these elements. By comparison, conventional analog circuits use the ratio of resistances to set the transfer functions of amplifiers, and the magnitudes of resistances to determine the operation of current-to-voltage and voltage-to-current converters. Finally, the values of RC products are used in active filters and signal generators to determine the frequency responses of those circuits. When one moves to the silicon chip and strives to achieve the same functionality in a much reduced area and using the tools of MOS technology, this is what one discovers. First, switches, small-value capacitors, and decent op-amps are easy enough to realize in MOS technology. Second, using that same technology, it is very difficult and wasteful of silicon die area to make resistors and capacitors with the values and accuracy encountered in audio and instrumentation applications [2,3]. As we will see in the subsequent sections, designers have overcome these difficulties by realizing that (1) resistors can be replaced by MOS switches that are rapidly turned on and off, and MOS capacitors, and that (2) the time constants arising from these simulated resistances and the MOS capacitors are given in the form of capacitance ratios. The fact that capacitor ratios control the time constants means that these constants now can take advantage of the superior matching of capacitances fabricated on silicon, as well as their ability to track each other with temperature. 

If these are undoubted advantages for the VLSI designer, what can the board-level circuit design expect to achieve with the use of switched-capacitors? For one thing, as we will see shortly, not only are the time constants of the switched-capacitor circuit superior in their control, but these time constants are tunable through the simple expedient of changing the frequency of the clock pulses that drive the circuit. Furthermore, the integrated circuit packages that are now available support a number of filtering functions in one package, thus reducing footprints needed on circuit boards to realize a given set of analog functions. 

Although switched-capacitors were developed in order to meet the need to incorporate analog, active filters on silicon along with digital functions, they have since found many other uses [2]. These include, besides filters, instrumentation amplifiers, voltage-to-frequency converters, data converters, programmable capacitor arrays, balanced modulators, peak detectors, and oscillators.

 

II. BASIC SWITCHED-CAPACITOR OPERATION 

The essence of the switched-capacitor is the use of capacitors and analog switches to perform the same function as a resistor.  This replacement resistor, along with op-amp based integrators, then forms an active filter.  Before delving too far into actual filter designs, however, it makes sense to ask why one would want to replace the resistor with such an apparently complex assembly of parts as switches and capacitors.  It would seem from the multiplication of parts that the switched-capacitor would be area intensive.  As a matter of fact, for the resistor values that one seeks in certain filter designs, this is not the case.  Furthermore, the use of the switched-capacitor will be seen to give frequency tunability to active filters.  Figure 1[2, 3] shows the basic setup for a switched-capacitor, including two N-channel Metal-Oxide Semiconductor Field-Effect Transistors (NMOS) and a capacitor.  There are two clock phases, , which are non-overlapping. The MOSFET's, either M1 or M2, will be turned ON when the gate voltage is high, and the equivalent resistance of the channel in that case will be low, . Conversely, when the gate voltage goes LOW, the channel resistance will look like .  With such a high ratio of OFF to ON resistances, each MOSFET can be taken for a switch. Furthermore, when the two MOSFET's are driven by non-overlapping clock signals, then M1 and M2 will conduct during alternate half-cycles.   

Figure 1.  Two NMOSFET's, driven by alternating, non-overlapping clock signals, comprise the basic switched capacitor network. 

This makes the two-MOSFET arrangement equivalent to a single-pole, double-throw switch (SPDT).  One can now use a symbolic switch picture, as in Figure 2a below, to represent the circuit.  The operation of this circuit is as follows. When the switch is thrown to the left, the capacitor will charge up to . When the switch is thrown to the right, the capacitor will discharge down to/charge up to . As a result of these consecutive switching events, there will be a net charge transfer of .  Now, if one flips the switch back and forth at a rate of  cycles/sec, then the charge transferred in one second is , which of course has the units of current.  One can claim that an average current, . If    is much higher than the frequency of the voltage waveforms, then the switching process can be taken to be essentially continuous, and the switched-capacitor can then be modeled as an equivalent resistance, as shown below in Figure 2b.  The value of the equivalent resistance is given by: 

                       (Eq. 1)

Therefore, this equivalent resistance, in conjunction with other capacitors, and Op-amp integrators, can be used to synthesize active filters.  It is now clear from Equation (1) how the use of the switched-capacitor leads to tunability in the active filters, by varying the clock frequency. 

                        (a)                                                                       (b)             

Figure 2. Equivalent resistor model for switched capacitor circuit in Fig. 1. 

This equivalent resistance has features which make it advantageous when realized in integrated-circuit form: 

(a)    High-value resistors can be implemented in very little silicon area. For example, a 1-MW resistor can be realized with a 10-pF capacitor switched at a clock rate of 100 kHz.

(b)   Very accurate time constants can be realized, because the time constant is proportional to the ratio of capacitances, and inversely proportional to the clock frequency:

. Capacitor ratios, especially in monolithic form, are very robust against changes in temperature, and clock frequencies can also be strictly controlled, so that accurate time constants are now available in the switched-capacitor technology. 

The principal constraint in using the switched-capacitor is that inherent in all sampled-data systems: the clock frequency must be much higher than the critical frequency set by the RC products in the circuit. Furthermore, on either side of the analog switches, i.e., the MOSFET's, there must be essentially zero-impedance nodes (voltage sources).  There are a number of other constraints which the unsuspecting designer/user might overlook [3, p. 725]: 

(a)    The equivalent resistance formed by the action of the switched-capacitor cannot be used to close the negative-feedback path in an op-amp all by itself.  One must recall that to ensure stability, the op-amp's feedback path must be closed continuously, while the switched-capacitor is a sampled-data construction of a resistor, and thus not continuous.

(b)   Circuit nodes cannot be left floating.  That is, there must always be a resistive path to ground so that charge does not build up on the capacitor plates.

(c)    The bottom plates of the MOS capacitors must be connected to ground or to a voltage source. There is an intrinsic, parasitic capacitance associated with the MOS capacitor's bottom plate [4]. This parasitic capacitance can be between 5% and 20% of the desired value; furthermore, it behaves nonlinearly with voltage [4].  Therefore, it must be connected to AC ground or a voltage source so that this nonlinear portion of the capacitance will not affect the overall response of the switched-capacitor filter. In practical terms, this means that capacitive voltage dividers with three or more capacitors, and circuits that switch both ends of a capacitor in sequence to the inputs of an op-amp, are used.

(d)   The noninverting pin of the op-amp should be kept at a constant voltage. If this pin is connected to the signal in some way, then the virtual short circuit between op-amp inputs means that the inverting input is no longer a virtual ground, and so an undesirable alteration of filter response due to the MOS capacitor's parasitic capacitance will occur (see item (c) above). 

 

III. SWITCHED CAPACITOR INTEGRATORS 

The op-amp integrator is the most frequently chosen building block for switched-capacitor filters.  The standard RC integrator is shown in Figure 3a, and its analysis and description can be found in any electronics text [5, 6]. 

 

 Figure 3a. Standard op-amp integrator, with transfer curve

   To review, the voltage transfer function of this circuit is given by

                 (Eq. 2)

where . Now, if one replaces the resistor by its switched-capacitor equivalent, as shown in Figure 3b, and use Eq. 1 as the resistor's value, one finds 

                 (Eq. 3) 

 

 

Figure 3b. Switched-capacitor integrator.

  Again, one notes the fact that this new integrator has no resistors, which take up excessive silicon die area. Also, the -3 dB frequency, , depends on a ratio of capacitances, not on an RC product.  The tolerances for ratios are much easier to control than the tolerances for products. Finally, this characteristic frequency of the integrator is inherently settable with a simple change in the clock frequency. 

The typical values of capacitances used in switched-capacitor technology range from 0.1 pF to 100 pF. These are low enough values that the stray capacitances of the MOS switches, of the interconnects, and of the "plates" of the switched-capacitors themselves can all have a significant effect on the desired frequency response of the filters designed with switched-capacitors. The effects of stray capacitance have been reduced greatly by dual-switch configurations [2, 7]. Figure 4 shows explicitly the clock phasing of the MOS switches which acts to eliminate the transient charge transfer through the stray capacitances, Cs1 and Cs2, also indicated in the figure.  In essence, charge transfer only takes place through the capacitor . Figures 5a and 5b show both the inverting and noninverting stray-insensitive integrator. The noninverting stray-insensitive integrator is obtained simply by switching the clock phasing on transistors M2 and M4.

  

Figure 4. Arrangement of extra MOSFET's and clock signals to make switched capacitor circuit insensitive to stray capacitances.

  

Figure 5a.  Switch setting used to realize stray-insensitive inverting integrator.

 

 

Figure 5b.  Switch setting for stray-insensitive non-inverting integrator.

 Because of the importance of the integrator to switched-capacitor filters, it is necessary to be familiar with the variants of the integrator.  These include the summing integrator, the differential integrator, the integrator/summer, and the lossy integrator. All of these play a role in the synthesis of switched-capacitor filters. The summing integrator, shown below in Figure 6, has a response given by: 

               (Eq. 4)

  

Figure 6.  Summing integrator.

  

Figure 7 shows the differential integrator.  The easiest way to understand this circuit is to look at what happens to the charge accumulation on the capacitor  when the switches are thrown to the left.  In this case, the capacitor charges up to a value of . When the switches are thrown to the right, the charge on the capacitor is poured into the op-amp's summing node. The average current, assuming the switching rate (= clock frequency) is high enough, is given by

     (Eq. 5)

 This results in a stray-insensitive output voltage of 

                      (Eq. 6)

 where  .

 

 Figure 7.  Differential integrator.

 The lossy integrator provides a simple, first-order lowpass response with gain. This circuit is realized by placing a switched-capacitor (i.e., a simulated resistor) in parallel with a feedback capacitor, Figure 8. In general, the easiest way to analyze the response of more complex switched-capacitor circuits such as this one is to replace all switched-capacitors with their resistor equivalents. Once the transfer function is found for a circuit with resistors (and discrete capacitors), then the switched-capacitor equivalents of the resistors (Eq. 1) can be placed back in the transfer function to obtain the final result. For the lossy integrator, the analysis proceeds as follows: 

          

where the "0" in  refers to the virtual ground at the op-amp's inverting input. The transfer function is obtained using these resistor equivalents: 

          (Eq. 7)

Now substituting the switched-capacitor equivalents for the resistors from Eq. 1, one obtains

 

                        (Eq. 8)

where   is the critical, or -3 dB, frequency of the lowpass filter.  Eq. 8 has the form of a lowpass filter multiplied by a gain proportional to the ratio of the two switched-capacitors.

  

Figure 8.  Lossy integrator, or first-order lowpass filter with gain.

  

IV. SWITCHED CAPACITOR BIQUADRATIC FILTERS

 The biquad configuration [8] normally features a lossy inverting integrator, a lossless inverting integrator, and a unity-gain-inverting amplifier.  In the standard active RC configuration, this requires three op-amps.  However, the switched-capacitor realization of the biquad needs only two op-amps to perform the same function. One op-amp performs the lossy inverting integration function, while the second op-amp performs lossless, noninverting integration. Although one can design an adequate switched-capacitor version of the biquad by making a resistor-by-resistor replacement in the standard RC biquad filter, such an implementation has been found to have unacceptably wide capacitance spreads, especially when higher filter Q's are sought [2]. Instead, Figure 9 shows the biquad filter with improved capacitance ratios. This circuit provides the highpass and bandpass responses. Just as with the analysis of the lossy integrator, a fairly complete analysis of this circuit will be made here. 

 


Figure 9.  Biquadratic bandpass and highpass filter provided by switched capacitor network, with improvements needed to reduce spread in capacitor values.

 The form of the bandpass filter function is at least clear: 

                     (Eq. 9) 

The use of a stray-insensitive switched-capacitor ( ) with alternating clock phases makes possible the noninverting form above. In order to complete the analysis, one has to calculate the highpass filter response.  The output node of the first op-amp, which gives the highpass filter response, can be seen to be the superposition of two signals at the summing node of the op-amp:                                                            

  (Eq. 10a)

 With   the expression in curly brackets simplifies, giving

(Eq. 10b)

 In order to obtain finally the bandpass transfer function, Eq. 10b is inserted into Eq. 9. The result of this manipulation is

                 (Eq. 11) 

Gathering like terms together, and multiplying both sides by , one obtains

 

              (Eq. 12)

 With , and isolating , the transfer function for the bandpass filter is finally found:

                          (Eq. 13)

 The corresponding expression for the highpass filter is simply obtained by inverting Eq. 9. Recalling that , one finds readily 

                 (Eq. 14)

 Although Eq. 13 does in fact display the standard form for the frequency response of a bandpass filter, there is nothing in the expression that gives the gain of the filter circuit.  In fact, from the form of the transfer function, and from simulation, it can be seen that the circuit in Figure 9 has no resonant gain. The same applies to the highpass filter expression in Eq. 14.  It is clear that a useful active filter circuit formed from switched-capacitors must possess some voltage gain in the passband. Of course, it is possible to simply "tack on" an amplifier to the output of the circuit. However, an elegant solution is to turn the second op-amp in Figure 9 into a summing integrator, in which the input signal is injected into the second op-amp via an equivalent resistance derived from another switched-capacitor. Figure 10 shows the implementation of this idea. In Figure 10, the SPDT switches, labeled S1, are shown closed in the first half of the clock cycle .  

The analysis of this circuit in order to derive the bandpass filter transfer function follows a path similar to the one followed in Eq. 13, although more tedious. We will simply present the main results here and proceed to the simulation of the circuit.  Superposition of input signals to each of the op-amp's input pins gives 

               (Eq. 15)

 After much manipulation, one finds that the transfer function for the bandpass filter is 

        (Eq. 16)

 In order to simulate the performance of switched-capacitor networks, it is necessary to realize that these circuits are a hybrid of both continuous and sampled-data signals.  In fact, switched-capacitor circuits are continuous in amplitude and discrete in time. Because of this combination, simulation with conventional circuit simulators such as PSPICEŌ presents a problem. The presence of switched voltages means that a transient analysis must be performed. At the same time, the desire to determine the frequency response across a wide range of frequencies means that a transient analysis must be performed for each desired frequency. This is a very time-consuming process because one has to wait until the steady state is reached. One way to overcome this problem within a SPICE-type simulator is to implement all the design equations in z-transforms. These z-domain models will allow one to perform frequency-domain simulation of complex switched-capacitor circuits. Although the z-transform is the rigorously correct mathematical tool for the analysis of sampled-data systems, it is not really available to the majority of engineering technology students. The model element used as the simulation kernel in SPICE for z-domain analysis is called the storistor [10]. It consists of conductances, a lossless transmission line for delay effects, and controlled sources. In order to simulate the frequency response of something as simple as an integrator, one is required to model as many as seven storistors, four capacitors, and an op-amp subcircuit. Given the long experience of many students and instructors with the SPICE program, this might be an acceptable alternative. However, the size of the input files for even simple switched-capacitor circuits (beyond the integrator), together with the mathematics required to understand the z-transform, will probably deter many from this approach. 

This paper will use a perhaps less well-known simulation package called APLACŌ [11-12] (originally Analysis Program for Linear Active Circuits). The APLAC program has been under continuous development since 1972. Since 1988, the Nokia Corporation, developers of wireless communication products, has sponsored continued improvement in the system design and electromagnetic capabilities of the APLAC program. The particular strengths of the APLAC program are its use of object oriented programming techniques, which permit easy adaptation of models to the circuit environment in which a component finds itself. Also, the program has a very extensive library of system level blocks, and the ability to model electromagnetic behavior of components. 

For our present purposes, one of the attractions of the APLAC program is its use of the convolution integral to model circuits with both frequency-dependent and transient behaviors. This feature is important for switched-capacitor circuits. The frequency-dependent portions of the circuit can be analyzed straightforwardly. However, the frequency response of the time-dependent portions of the circuit (such as switches and sources) is calculated by creating a frequency-domain equivalent circuit by means of the convolution integral. 

In the circuit of Figure 10, the following values of components and parameters are used: 

These values give a resonant (center frequency) voltage gain of A0 = 10, a Q = 50, and a center frequency of  f0 = 20 kHz. The results of the simulation are shown below, in Figure 11.

 

 Figure 10. Biquadratic switched-capacitor filter with gain setting from summing integrator. The switches S1 are closed on the first half-cycle of the clock waveform. 

 

Figure 11. APLAC simulation of bandpass filter with gain, from Figure 10. From Probe tool of simulator, peak gain is 3.32 dB, and center frequency is at f0 = 17.8 kHz.

The results show a simulated gain of only 3.3 dB, and the center frequency is off by ~2 kHz. This inaccuracy is due to the use of ideal MOS switches in the hand calculations leading up to Eq.  16. The APLAC model for the switches assumes an Ron = 100 W, and an Roff  = 100 kW.  APLAC has the capability to optimize the circuit's component values to achieve desired circuit behaviors.

  

V. APPLICATION OF MF-10 

The MF-10 is a universal switched-capacitor filter supplied by National Semiconductor [1]. The MF-10 uses the two-integrator loop structure to realize lowpass, highpass, bandpass, notch, and allpass functions through externally chosen, discrete, resistors. The actual switched-capacitor integrators are internal to the chip, while the external resistors give the user flexibility in configuring his/her own response. However, to take advantage of component tracking with temperature, etc., all responses are designed to be functions of resistor ratios only.  Figure 12 shows the summing amplifier and two-integrator cascade internal to each section of an MF-10. The tunability of a particular filter's critical frequency, , is determined by a logic level applied to a 50/100/CL frequency ratio programming pin. In other words, the critical frequency will be 

; if the programming pin is tied to ground, the divisor is 100, otherwise, if tied to a HIGH (positive power supply), the divisor will be 50. Figure 12 shows how the notch, bandpass, and lowpass filter functions are realized by the MF-10. Because the summing amplifier is outside the two-integrator loop, this configuration will be faster and allow a greater range of operating frequencies. 

 

Figure 12. Two-integrator loop, with external resistors, used in MF-10 to realize notch, lowpass, and bandpass filters. 

The analysis of the transfer functions for the three transfer functions mentioned above follows the pattern in Eq. 13-14. By inspection, one sees that 

                                     (Eq. 17a)

                                    (Eq. 17b)

                                                  (Eq. 17c)

 

Eliminating  to obtain , the transfer function for the notch filter is obtained: 

          (Eq. 18)

Similar manipulations give the bandpass and lowpass responses. The bandpass response is:

              (Eq. 19) 

The lowpass response is:

              (Eq. 20)

 

VI.   SWITCHED-CAPACITOR IMPLEMENTATION APPLIFIER ¾ LTC1043 

Although the initial impetus for the development of the switched-capacitor was the opportunity and need to synthesize active filters that would be compatible with MOSFET technology, the early 1980's found many other uses for the switched-capacitor. Linear Technology has developed the LTC1043 [9], which contains dual switched capacitor networks, along with an on-chip non-overlapping clock generator, oscillator, and charge balancing circuitry. The clock generator controls both of the switch networks, while the charge balancing circuitry is designed to cancel any effects due to stray capacitance. The on-chip oscillator has a fixed frequency of 185 kHz. An external capacitor can be connected across pins 16 and 17 (for the instrumentation amplifier) to yield any desired clock rate. The desired clock rate can be found from 

; the 24-picofarad capacitance is the internal capacitance responsible for the oscillator’s fixed frequency.  

Among the circuits developed from the LTC1043 are instrumentation amplifiers, lock-in amplifiers for detecting extremely small parameter shifts in sensor applications, and signal conditioners for platinum resistance temperature detectors (RTD), relative humidity sensors, and LVDT’s. The instrumentation amplifier is a standard op-amp circuit presented in many electronics texts [5-6], and is designed to amplify small difference signals such as might be found in measurement or transducer applications. At the same time, common-mode or noise signals picked up by the lines feeding the amplifier must be suppressed, especially as these signal levels are often larger in amplitude than the sought-for difference signals. Figure 13 shows the LTC1043 combined with a standard non-inverting op-amp to give an instrumentation amplifier with a common-mode rejection ratio (CMRR) of  >120 dB. Figure 14 shows the same circuit with the ½ LTC1043 as a black box.


Figure 13. Instrumentation amplifier using ½ of LTC 1043 switched-capacitor, along with LF356/353 op-amp in non-inverting configuration.

 
Figure 14. Switched-capacitor-based instrumentation amplifier, with ½ LTC1043 shown as a black box. The pin numbers in Figure 13 are the pins in the black box in this figure.

 The operation of this circuit is as follows. First, the dual switch, when flipped to the left, charges the capacitor C1 up to the difference V1 – V2. Second, on the next clock pulse, the switches will then dump the charge represented by that voltage difference onto C2. Third, the continuous clocking from the oscillator will force C2 to eventually develop a voltage equal to the difference voltage. Finally, the difference voltage, with the common-mode signal stripped off by the LTC1043 is amplified by the op-amp. It is interesting to observe several features of this circuit and compare them to the standard instrumentation amplifier. By using the capacitor C1 (the so-called “flying capacitor”), the common-mode voltage present at the inputs is looking into a capacitive voltage divider, between the C1 and the LTC1043’s parasitic capacitance. This parasitic capacitance is typically less than 1 picofarad, so the AC value of the CMRR is > 120 dB. By comparison, Analog Device’s AD624 instrumentation amplifier can go as high as 130 dB for high gains, up to 60 Hz. Because of the capacitive voltage divider from the LTC1043, this instrumentation amplifier shows higher CMRR, over a wider range of voltage gains, and to a higher frequency.  

VII.  CONCLUSIONS 

This paper has presented the essentials of operation  of switched-capacitor networks, with a special emphasis on its use in designing active filters. Unlike active filters based on the conventional op-amp, switched-capacitor filters have critical frequencies that are easily pin-settable. Furthermore, they require less power than the conventional op-amp based network because of their reliance on CMOS technology. Finally, for the functionality provided on a single chip, they take up less room on circuit boards. Alternate use of the switched-capacitor network in an instrumentation amplifier has also been presented. The operation of this device is a little easier to digest for some students than discussion of active filters; it is hoped that instructors and students can use the information herein to extend their acquaintance with modern integrated circuits. 

The presentation of results here is in a form which instructors and upper-level students in electronics technology can adapt to the curriculum in technology programs. 

REFERENCES 

1.      Introducing the MF-10: A Versatile Monolithic Active Filter Building Block, by Tim Regan. National Semiconductor Application Note 307, August 1982.

2.      Design with Operational Amplifiers and Analog Integrated Circuits, by Sergio Franco. McGraw-Hill Book Company, New York, 1988. Chap. 13.

3.      Bipolar and MOS Analog Integrated Circuit Design, by A.B. Grebene. Wiley-Interscience, John Wiley & Sons, New York, 1984. Chap. 13, pp. 703-752.

4.      Fundamentals of MOS Digital Integrated Circuits, by J.P. Uyemura. Addison-Wesley, Reading, MA, 1988. Chap. 8.

5.      Electronic Devices, 5th Edition, by T. L. Floyd. Prentice-Hall, Englewood Cliffs, NJ, 1997. Chap. 14.

6.      Microelectronic circuits, 3rd Edition, by A. Sedra and K.C. Smith. Saunders College Publishing/HRW, Philadelphia, PA, 1991. Chap. 2.

7.      "Stray Capacitance Insensitive Switched Capacitor Filters," M. Hasler. Proceedings of the IEEE International Symposium on Circuits and Systems, 1981.

8.      Analog Filter Design, by M.E. Van Valkenburg.  Holt, Rinehart, and Winston, New York, 1982.  Chap. 5.

9.      Applications for a Switched-Capacitor Instrumentation Building Block, by Jim Williams. Linear Technology Application Note 3, July 1985.

10.  CMOS Analog Circuit Design, 2nd Edition, by P. Allen and D. Holberg. Saunders College Publishing/HRW, Philadelphia, PA, 1998. Chap. 9.

11.  APLACŌ 7.0 User's Manual, Helsinki University of Technology, Circuit Theory Laboratory & Nokia Corporation Research Center, 1998. Available at http://www.aplac.hut.fi/

12.  "Fast Analysis of Nonideal Switched-Capacitor Circuits using Convolution," H. Jokinen, M. Valtonen, and T. Veijola. 11th European Conference on Circuit Theory and Design, Davos, Switzerland, 1993, pp.941-946.

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